1. Field of the Invention
The present invention relates to a high-frequency line-waveguide converter for converting a high-frequency line which is used in a microwave or millimeter wave band and constitutes a high-frequency circuit (such as a strip line, a microstrip line, a coplanar line, or a coplanar line having a grounding conductor) to a waveguide, the high-frequency line-waveguide converter having succeeded in facilitating system mounting by providing connection between a high-frequency circuit and an antenna, or between high-frequency circuits, through a waveguide.
2. Description of the Related Art
In recent years, as we move into the so-called advanced information age, signals in an increasingly wider range of frequencies have been sought after for information transmission. Consequently, studies have been conducted as to the use of signals falling in a range from micro-wave-band frequencies (1 to 30 GHz) to millimeter-wave-band frequencies (30 to 300 GHz). For example, versatile systems employing high-frequency signals in millimeter-wave-band frequencies, such as a vehicular radar, have been proposed to date.
However, such a high-frequency system poses the following problem. In its high-frequency line constituting a circuit, high-frequency signals are significantly attenuated due to the high frequency level. For example, in a case where the high-frequency line is built as a micro-strip line, a dielectric loss as observed in the dielectric substrate is increased proportionately with a frequency (assuming that the dielectric loss tangent is independent of the frequency), and also a conductor loss as observed in the line conductor is increased proportionately with the square root of the frequency. As is understood from this fact, even in the micro-strip line, when the usable frequency is as high as 1 GHz to 10 GHz, the dielectric loss becomes 10 times higher, and the conductor loss becomes approximately 3.2 times higher. To compensate for these losses, it is necessary to make heavy use of expensive high-frequency components that are lower in noise but higher in efficiency and gain, resulting in the system as a whole being expensive.
It has been known that, as compared to such a high-frequency line having a micro-strip line structure, a waveguide incurs less transmission loss of high-frequency signals. For example, in a waveguide WR-28 (EIA (Electronic Industries Alliance) standard) which is operated at a frequency band of 26 to 40 GHz, a loss is expected to be approximately 0.005 dB/cm at 40 GHz. This figure is far smaller than a value of a loss as observed in a micro-strip line composed of an alumina ceramic substrate: approximately 1 dB/cm. These are the following reasons for this. As compared to an ordinary high-frequency line (generally designed with an impedance of 50Ω), a waveguide has a greater impedance (generally designed with an impedance of the order of 500Ω, though it varies with frequencies). In the waveguide, the effect of electric field energy propagating through a dielectric substance is more significant relative to that of signal energy to be transmitted, and, used as the dielectric substance is air having a dielectric tangent of substantially nil. Moreover, a current flowing through the tubular wall of the waveguide, which gives rise to relatively weak magnetic energy, can be kept low. Further, since the current flows through a relatively wide area in the tubular wall of the waveguide, the resultant electrical resistance decreases, resulting in the conductor loss being reduced.
Besides, two or more waveguides are generally coupled to each other by a screw. Therefore, their attachment and detachment can be made with ease. For example, in a case where a waveguide is used to provide connection between a high-frequency circuit module and an antenna, examination is conducted, before assembly, on each of the components using a waveguide port, so that an RF front end is assembled using a combination of conforming components. As a result, the manufacturing yield can be enhanced. With this being the situation, in most of conventional front ends, a waveguide has been used to achieve transmission between a high-frequency circuit module and an antenna, in consideration of its long transmission distance.
FIG. 9 is a sectional view for explaining the structure of the RF front end. According to FIG. 9, a front end 60 is constructed by connecting a module 61 to an antenna 62 via a waveguide member 63. The module 61 is mounted on a metal chassis 65 having a waveguide opening 64. Moreover, the front end 60 includes a micro-strip substrate 66 and a high-frequency line-waveguide converter 68. In the micro-strip substrate 66 is formed a micro-strip line acting as a high-frequency line. The high-frequency line-waveguide converter 68 is composed of the waveguide opening portion 64 and a waveguide composed of a short-circuit terminating member 67. Connected to the micro-strip line of the micro-strip substrate 66 by wire bonding is a wiring board 69 having high-frequency components mounted therein.
In the front end 60, the high-frequency line-waveguide converter 68 is constructed as follows. At a distance of ¼ times of a signal wavelength of a high-frequency signal from the short-circuit terminating face of the short-circuit terminating member 67, a probe formed on the micro-strip substrate 66 (the portion in which a line conductor is extended but no grounding conductor is formed) is inserted, from the side face of the waveguide, by a length of substantially ¼ times of a signal wavelength. In the waveguide, this probe functions as an antenna for radiating a high-frequency signal as an electromagnetic wave into the waveguide. Half of the electromagnetic wave radiated into the waveguide is directly transmitted to the waveguide member 63 located below, while the other half is transmitted toward the short-circuit terminating member 67 located above. The electromagnetic wave transmitted toward the short-circuit terminating member 67 is phase-inverted at the short-circuit terminating face so as to be totally reflected. The totally reflected electromagnetic wave returns to the probe portion and is then combined with an electromagnetic wave which is directly radiated downwardly from the probe. At this time, by adjusting the distance between the probe and the short-circuit terminating face to ¼ times of a signal wavelength of a high-frequency signal, the length of the double optical path (the probe—the short-circuit terminating face—the probe) is one-half wavelength long. Thus, the electromagnetic wave reflected from the short-circuit terminating face and the electromagnetic wave radiated directly from the probe are in phase opposition due to the optical-path difference. Eventually, the electromagnetic wave reflected from the short-circuit terminating face is phase-inverted when reflected from the short-circuit terminating face, and is then further phase-inverted due to the optical-path difference. Consequently, the electromagnetic wave reflected from the short-circuit terminating face is identical in phase with the electromagnetic wave directly radiated downwardly from the probe, and is then transmitted to the waveguide member 63 located below.
At this time, in order for the probe to function as an antenna, the length of its part inserted into the waveguide needs to be exactly ¼ times of a signal wavelength. Moreover, in order for the electromagnetic wave which has been radiated upwardly from the probe and reflected from the short-circuit terminating face to be identical in phase with the electromagnetic wave radiated downwardly from the probe, the distance between the probe and the short-circuit terminating face needs to be exactly ¼ times of a signal wavelength. Hence, the characteristics are significantly varied depending on the position of the micro-strip substrate 66 acting as an antenna or the height of the short-circuit terminating member 67.
The high-frequency line-waveguide converter 68 is formed on the metal chassis 65, together with the wiring board 69, by assembly. Therefore, when the high-frequency line-waveguide converter incurs a significant conversion loss due to positional deviation of each component, all of the constituent components become useless. This gives rise to a problem of the assembly yield being low.
To solve such a problem, for example, Japanese Unexamined Patent Publication JP-A 6-112708 (1994) discloses a waveguide-planer line converter which is constructed by forming, on a dielectric substrate, a grounding conductor acting as a short-circuit terminal of a waveguide and a radiating conductor acting as an antenna. In the construction disclosed in JP-A 6-112708, the distance between the short-circuit terminal of the waveguide and the radiating conductor is adjusted to ¼ times of a signal wavelength of a high-frequency signal. This structure is the same as that of conventional converters.
However, this construction poses the following problems. The dielectric substrate for adjusting the distance from the short-circuit terminal of the waveguide to the radiating conductor and the dielectric substrate for constituting a micro-strip line are identical, and the thickness of the dielectric substrate needs to be ¼ times a signal wavelength. This requires impedance adjustment for the micro-strip line be conducted solely by changing the conductor width of the line conductor. Thus, to set the impedance of the micro-strip line at a certain value, when the substrate is made thick, the conductor width of the line conductor needs to be increased, whereas when the substrate is made thin, the conductor width needs to be decreased. Consequently, depending on signal frequencies, a difference in conductor width is caused between the line conductor and its counterpart line conductor of the micro-strip line that are connected to each other, resulting in occurrence of signal reflection.
Moreover, since this converter is so designed that the thickness of the dielectric substrate is adjusted to ¼ times of a signal wavelength, when the signal frequency is low, the thickness of the dielectric substrate is increased, whereas when the signal frequency is high, the thickness is decreased. Thus, when the signal frequency is high, the dielectric substrate is so thin that its strength is decreased. For example, in a case where the signal frequency is 76 GHz and the relative dielectric constant of the dielectric substrate is 9, the thickness of the dielectric substrate is given as approximately 0.33 mm. In this case, depending on materials used for the substrate, there is a possibility that the dielectric substrate is deformed or broken.
Further, in this converter, part of the radiating conductor that functions as an antenna is inserted into the waveguide. Thus, if the waveguide-planer line converter undergoes positional deviation and this causes a change in the length of the part functioning as an antenna, the antenna characteristics are varied, resulting in degradation of the conversion efficiency of the waveguide-planer line converter.
In addition, for proper use of this converter, a concavity needs to be provided in part of a waveguide opening formed in a metal-made casing, to prevent occurrence of short-circuiting in the radiating conductor. This means that a portion which should act as the tubular wall of the waveguide is partly missing. Thus, an electromagnetic wave leaks from this concavity, resulting in degradation of the conversion efficiency. Besides, since formation of such a concavity is absolutely necessary, the waveguide and the casing need to be subjected to extra processing.
To solve such problems, for example, a microstrip-waveguide converter is disclosed in International Patent Application published under the PCT as International Publication No. WO96/27913 (1996). In this construction, a micro-strip line is formed on a top surface of a dielectric substrate, and a slot, acting as an antenna, is formed on a grounding conductor layer of a bottom surface of the substrate. This micro-strip-waveguide converter disclosed in WO96/27913 is so designed that the thickness of the dielectric substance, from the slot to the waveguide, is adjusted to ¼ times of a signal wavelength of a high-frequency signal. In this case, the difference in impedance between the slot and the waveguide is compensated for by a quarter-wave matching device constituted by a dielectric substance.
According to this construction, an electromagnetic wave emitted from the slot is reflected from an interface between the dielectric-substance matching device and the waveguide, and is then reflected from the grounding conductor layer carrying the slot, and eventually returns to the interface between the matching device and the waveguide. At this time, by adjusting the thickness of the matching device to ¼ times of a signal wavelength, the difference in optical path between the electromagnetic wave reflected from the interface and returned (reflected wave) and the electromagnetic wave transmitted directly to the interface from the slot (direct wave) becomes one-half signal wavelength long. Since the reflected wave is phase-inverted when reflected from the grounding conductor layer, at the interface, the direct and reflected waves are in phase and mutually intensified, and are then transmitted to the waveguide.
According to this conversion structure, the conversion characteristics are varied greatly with the thickness of the matching device. However, in this case, since the matching device is formed integrally in the dielectric substrate, variation in the thickness of the dielectric substance can be minimized. Accordingly, the variation of the conversion characteristics can be suppressed. Moreover, by covering the micro-strip side of the dielectric substrate with a cap, not only it is possible to achieve conversion to a waveguide, but it is also possible to achieve hermetic sealing of the micro-strip side.
In this structure, the high-frequency line and the slot are coupled to each other by exploiting electromagnetic coupling between different layers. This electromagnetic coupling, together with the above-described matching device, plays a key role in conversion action. However, the characteristics of the electromagnetic coupling are varied with the dimension of the slot and the length of the stub (part of the high-frequency line jutting from the slot), that is, varied with the relative positional relationship between the high-frequency line and the slot. Hence, in this structure, the conversion characteristics are varied greatly with the dimension of the slot and the length of the stub. Since the high-frequency line and the slot are arranged on different layers, the length of the stub, which is determined in accordance with the relative positional relationship therebetween, is likely to change. This gives rise to a problem of the conversion characteristics being varied easily.
Moreover, in this structure, since the slot is arranged in the dielectric substrate, neither the length and width of the slot nor the length of the stub can be examined externally with ease. This makes it difficult to stabilize the characteristics by conducting an examination.